Performance analysis of ZVS and ZCS boost converters for Solar Photo Voltaic Applications

. This paper describes a useful modulated, asymmetric full-bridge converter for sustainable solar PV applications. The suggested converter achieves zero-voltage switching (ZVS) on the main side's power switches and reduces circulating current loss by utilizing a full-bridge architecture and an asymmetric control method. The blocking capacitor and the leakage inductance of the transformer operate in resonance to further provide the Zero Current Switching (ZCS) off for the output diode without the need for any further circuitry. Output diode reverse recovery is no longer a concern as a result. Additional connections to the input voltage include the power switch voltage stresses. Because of these features, the suggested converter has the infrastructure to reduce power losses. The buildings of renewable energy sources can benefit especially from it


Introduction
Due to global capital shortages and environmental degradation, research on alternative energy sources, including fuel and solar cells, has been gradually progressing in the industrial sector [1].Renewable energy sources are generally used to produce low-voltage power.PV cells, which depend on their surroundings, provide varying low-voltage energy.A front-end converter is therefore needed to transfer power from a low-voltage source to a high-voltage load.Usually, the power capability of the front-end converters is no more than 250 W. The front-end converter's power capability may increase as cell technology develops.A bigger power capacity can lower the cost per watt as well.Thus, the increased power rating of the advanced cells must be accommodated by the enhanced front-end converter power rating to reduce the per-watt costs of renewable energy systems [2].Front-end converters, forwardflyback converters with an active clamp and voltage doubler, LLC converters, and full-Corresponding author: gireesh218@gmail.com* * , 010 (2023) E3S Web of Conferences ICMPC 2023 https://doi.org/10.1051/e3sconf/20234300100404 430 bridging phase-shift (PSFB) converters are the topology that frequently comes into play for power capacities [3][4].The switches were constructed effectively by magnetizing inductance, parasite capacitance, and an active clamping circuit known as Zero Voltage Switching (ZVS).The Zero Current Switching (ZCS) of transformer secondary side diodes, which is induced by the resonant current produced by the leaky inductance and the resonant capacitor, is provided by a forward/flyback converter with an active-clamp voltage doubler.The voltage control over the transformer's primary switches that forward/flyback converters have is far larger than the input voltage.Low resistance RDS (on) should also not be used with the MOSFET [5].An LLC Resonant Converter can utilize any application requiring changing input and output voltages, high-performance standards, high power density, and diverse frequency management.However, due to the extremely wide bandwidth, the frequency must be raised significantly to provide a sufficient regulated voltage.Regular LLC resonance architecture as the front-end converter is very important to install due to its challenge of utility above variable input voltage under various load situations [6][7][8].For high performance, full-bridge phase shift transformers (PSFB) are widely utilized in medium-strength applications.Due to its straightforward design, switches operate with gentle switching without the need for additional components [9][10][11][12].Even so, the full-scale input conversion unit with the phase-shift control mechanism is inappropriate for use with fluctuating input voltage due to some significant flaws in its monitoring system, including a small number of ZVS sets of lagging leg switches, a failure of the duty cycle, a significant failure of the current, and voltage variations around the output diode.The enormous voltage surge is potentially harmful, especially for high-voltage applications [13].Freewheeling time is required to tackle the problem of restricted ZVS stretch with variable input voltage.Therefore, a big flowing current increases the main portion's conduction loss.The trailing leg shifts also result in higher switching losses.Furthermore, ZVS operation of trailing leg switches under light loads cannot be guaranteed due to insufficient energy stored in the leaky inductor.Additional tools are thus frequently used to address ZVS functioning issues.However, using such devices to expand ZVS may result in an increase in lead losses and a reduction in the converter duty cycle [14].Due to the resonance of the leak inductance with the power condenser, the PSFB converter [15] provides a variety of ZVS for rear leg switches and lowers the flow during freewheeling.However, the tiny resonance frequency could cause extreme stress.With the addition of the clamping circuit, the conventional PSFB converter is thus also not intended for varying input voltage.Traditional PSFB converters, as was previously said, are unable to operate with high efficiency when the input voltage is oscillating due to the convoluted circuit design and power losses [16][17][18].
The complete bridge converter in this work employs part resonance and the fixed frequency controller APWM, allowing ZVS to turn on the switches and ZCS to turn on the output diode [19][20][21].The output diode voltage stress is kept to a minimum even in the absence of voltage spikes, and the stresses on the main side of the power switch (S1-S4) are coupled to the input voltage.The unbalanced output components and the secondary side of the transformer's reverse recovery issue burden the APWM control system [22][23][24].On the other hand, the ZCS switch-off in the proposed APWM full-bridge converter can solve these issues.Because the freewheeling period is not more reliable than conventional front-end converters under varying input voltages, the circulating current loss at the transformer's primary sides is removed.Therefore, the recommended asymmetric boost converter is better suitable for highefficiency demands with quickly varying input voltage.

Steady State operation of the converter 2.1 Operating modes of the proposed converter
A high-performance asymmetric boost converter architecture with a constrained input voltage range is shown in Figure 1(a).Except for the dc blocking and the secondary side of the transition, the architecture of the recommended conversion device is comparable to that of a traditional full-bridge converter.The operating waveforms of the suggested inverter are shown in Fig. 1(b).The functionality of the suggested converter is split into six modes during a transition time Ts.
Mode 1: S2 and S3 switches periodically turn off.The output capacitors CS1 and CS4 across switches S1 and S4 will be discharged by the primary current ip, and the capacitors CS2 and CS3 across switches S2 and S3 will start charging.Due to the tiny output capacity of the Coss switch, this mode has very brief intervals and is inconsequential.The magnetic current (Im) and primary current (IP) are thought to have constant values.

Mode 2:
The voltages VS1 and VS4 of the switches are zero at time t1.The negative current flows through the DS1 and DS4 diodes while S1 and S4 switches are triggered.ZVS then operates on the n-turn of switches S1 and S4, with resonance occurring between the primary Lm+Llk inductor of the transformer and the dc condenser Cb.However, noise is present since the resonant duration is significantly greater than one Ts.Regardless of how the dc blocking

Analysis of the proposed converter
The input voltages Vd and Vb of the dc-blocking condenser Cb average voltage constitutes an inductor on the main side of the n-transformer if the S1 and S4 switches operate at duty rate D. When the S2 and S3 switches are in 1-D service ratio, the output diode Do is activated and the main side inductor of the transformer is applied with the reflected output voltage Vo/n.The primary current ip equations in ( 1) and ( 2) may be determined because the resonant period of the resonant network is significantly longer than the dead-time period.As Llk's leakage induction is far smaller than Lm's magnetizing inductance, the Llk leakage is marginal on the primary side in Mode 4 and Mode 5.The following equation is also possible: Through ( 8)-( 10), the voltage gain is written as follows between Vd and Vo input voltage: Since the leakage inductance is irrelevant, ( 10) and ( 11) are reflected in the average voltage Vb of the dc condenser.
Because of the charge balance of the dc blocking condenser Cb, the average primary current Ip value in steady state is 0. Consequently, the magnetic current Im and average output current Io values fit the following definition.
Of the frame, the typical magnetization current Im is also possible.
The new IP(t1) and ip(t3) are stated by ( 1), (13), and ( 14) The resonant current ( 6) with ( 13)-( 16) can be seen For the S1 and S4 turn-off, the key current ip(t1) should be negative before S1 and S4 are switched on.Therefore, the following can be derived from ( 15) the ZVS state.
The min-the-max theorem is structured as Where Ro, min = Vo/Io, max is the minimum output resistance and Io, max is the maximum output current.Dmax is the total service ratio for the S1, S4, and minimum input voltage Vd, min switches.One definition of Dmax is from (11).
Due to the variations in input voltage Vd and turn ratio, the duty ratio of the switches S1 and S4 can be seen in Fig. 5.To meet the ZVS requirement, the magnetizing inductance Lm from (19) and ( 20) should be built as follows.
The essential magnetizing inductance value Lm, where Ts is the switching time to meet the ZVS turn-on state of the switches based on the fluctuation of the service ratio D. The ZVS turn-on states of the S1 and S4 switches may be described similarly to the S2 and S3 switches.This shows that the ZVS function for S2 and S3 is possible when the main current ip(t3) is positive.The ZVS condition S2 and S3 is expressed as follows, according to (16): Irrespective of the load differences, the left-side words of ( 22) are still positive.Therefore, the ZVS function of the S2 and S3 switches can still be accomplished.
There must be enough time between two switch pairs for each ZVS turn-on operation so that the voltage may completely drain across the switch's Coss output capacitance.The minimal dead time for the deceased may be calculated as, since IP(t1)=im(t1) is taken to be a constant value during the dead period.
Typically, the primary current ip(t3) is bigger from ( 15) to ( 16) than the main current ip(t1)'s absolute value.Therefore, ( 23) may be reduced to The main current ip(t1) must be negative for ZVS functioning.As a result, ( 24) can be stated as seen in, In the practical configuration of the magnetizing inductance, attention should be given to the minimum dead time, i.e. dead, since, i.e. dead is often less than (1−Dmax) Ts.

ZCS switching
The resonant angular frequency ubiquitous should be larger than the critical one to accomplish the ZCS turn-off condition of the output diode Do.The critical angular frequency may be determined as follows since the critical state is ip(Ts)=im(Ts) in the range of = 2Ts = 0 and D = Dmax: The minimum turn-on time of the S2 and S3 switches is given by tS2, min.In general, the magnetizing inductance Lm is designed to have a small negative value for the magnetizing current im(t1) to reduce the converter's conductive loss.(26) The following may be deduced from this statement, It is articulated in (21), (27), as seen in As a result, the crucial angular frequency may be calculated using the method described in The following (29) relationship must be satisfied by the dc blocking capacitance Cb: The significant resonant capacitance Cb satisfies the output diode's ZCS turn-off condition and accepts the variations in the duty cycle D.

Results & Discussions
Both test parameters are suitably configured to reach the highly efficient range with low input voltage.Application requirements of the proposed APWM full-bridge converter are addressed in this section with soft switching techniques for their high-performance operation, and theoretical waveforms reflect soft switching of power switches and output diodes.Furthermore, the calculated power efficiency could be seen according to the input voltage and the output power.The APWM full-bridge converter's output voltage and maximum power are Vo=350V and Po=400W (Ro, min=306).The duty ratio is the optimal Dmax task to accommodate the full output power at the lowest input voltage.As well as the transformer's turn ratio is chosen as n=8 (Np: Ns=6:48) in (20).The magnetizing inductance Lm should be less than 43μH from the ZVS turn-on condition (21) of the switches S1 and S4 to guarantee the ZVS operation of the control switches.The higher inductance of magnetization causes the lower root mean square values of the main and secondary current, decreasing the conduction loss.However, transformer saturation should be considered in the frequency of operation.The magnetizing inductance Lm is therefore essentially chosen as 28μH, and the inductance of leakage Llk is calculated as 0.45μH.You can pick Cb= 7.6μF.

Conclusions
For renewable energy conversion systems where the input voltage may fluctuate significantly between 40V and 80V, this research investigates the APWM full-bridge converter.The output diode runs on ZCS without additional modules, whereas both power switches run on ZVS.Both power switches also clamp the input voltage.To avoid power losses, the suggested converter has the necessary structure.These benefits make the suggested converter suitable for adjusting input voltage on renewable energy switching systems.An APWM full-bridge converter implementation is given to demonstrate the suggested idea.50V input voltage and 400W output power have a 97.83% overall efficiency.

, 3 :
010 (2023) E3S Web of Conferences ICMPC 2023 https://doi.org/10.1051/e3sconf/20234300100404 430 capacitor Cb and the input stresses change, the main current's path changes and is sustained relatively linearly as follows: Here, Vb = Average voltage and Vd = Reference input voltage Mode The switches S2 and S3 turn off at times too.The output capacitors CS1 and CS4 across the switches S1 and S4 will get discharges through the primary current ip, and the capacitors CS2 and CS3 across the switches S2 and S3 begin charged like mode.1, here the primary current ip and magnetizing current Im are fixed values.

Mode 4 :Mode 5 :Mode 6 :
Similar to mode 2, the S2 and S3 switches are flipped ZVS at t3.The transformer's secondary side receives the energy stored in the magnetizing inductance, and the voltage across it is clamped with a reflected output voltage as, Where n = Np / Ns.Considering that the new output io expresses the difference between the main current ip and the magnetizing current Im, The resonance is between the dc condenser Cb and the transformer leakage llk.The voltages across the primary-side leak inductor Llk range from Vd+Vb to the observed secondary-side output voltage Vo/n.The state equations can therefore be written accordingly, By solving the equations (4) & (5), The primary current ip can be written as, Here, we = angular frequency at resonance and Zr = resonant impedance and can be written as, At time t4, the ip current flow reverses and remains zero.The magnetizing current typically switches axes at this period.The output current io is zero after this mode with resonance.This mode expires when the output current io is 0. The current output io becomes empty at t5, signaling the end of the resonance that started in mode 4. The Do output diode is left on until the S2 and S3 switches are turned off.The magnetizing current in this mode and the key current ip are the same.The output diode Do is thus turned off by ZCS.